Conventional high power amplifiers such as those utilized for telecommunications purposes, including for example Doherty power amplifiers, often have bias line implementations so as to achieve increased baseband equivalence resonance. Such implementations can for example involve the use of dual symmetrical bias lines (or feeds) that achieve increased baseband equivalent resonance by a factor of the square root of two. Such an improvement can arise from the parallel combination of both bias line equivalent circuits (assuming both lines are symmetrical and have the same dimensions). This in turn will reduce (by half) the equivalent inductance (e.g., in nanoHenries) from the bias lines presented at the package reference plane.
Further for example, FIG. 1 shows a circuit diagram 100 intended to illustrate an example high power amplifier (or package) 102 coupled to first and second bias lines 104 and 106, respectively. As shown, the high power amplifier 102 includes a power transistor 108 that in the present example is a LDMOS (Laterally Diffused Metal Oxide Semiconductor) transistor having a drain that is coupled to each of a first inductor 110 and a second inductor 112. The second inductor 112 is coupled in series between the drain of the power transistor 108 and a capacitor 114, and the capacitor 114 is coupled in series between the second inductor 112 and an additional port 116, which can be ground. The first inductor 110 is coupled between the drain of the power transistor 108 and an output port 118 of the high power amplifier 102. As for the first and second bias lines 104, 106, each of those bias lines includes a respective transmission line 120 and a respective set of first, second, and third capacitors 122, 124, and 126. The respective transmission line 120 of each of the bias lines 104, 106 is coupled between the output port 118 of the high power amplifier 102 and each of the three capacitors 122, 124, and 126 of the respective bias line. All three of the respective capacitors 122, 124, and 126 of each respective bias line 104, 106 are coupled in parallel with one another between the respective transmission line 120 and another port, which can be the additional port 116 (and can be ground). Further as illustrated, each of the bias lines 104, 106 can be represented by a small signal equivalent circuit 128 including an inductor 129 and a capacitor 130 coupled in series with one another (e.g., between two ends of the equivalent circuit that are each tied to the additional port 116 and can be grounded).
Given such an arrangement, it should be appreciated that a first resonant frequency fR1 of the first bias line 104 and a second resonant frequency fR2 of the second bias line 106 can be represented, respectively, by the following Equation (1) and Equation (2):fR1=1/(2π(LtotCtot)1/2)  (1)fR2=1/(2π(LtotCtot)1/2)  (2)Given this to be the case, the overall increase in the baseband equivalent resonance that arises from the combination of the first and second bias lines 104, 106 can be determined as the first resonant frequency fR1 divided by the second resonant frequency fR2, which as shown by the following Equation (3) has a value of the square root of two.
                                                        f                              R                1                                      =                          1                              2                ⁢                                                                  ⁢                π                ⁢                                                                            L                      tot                                        ⁢                                          C                      tot                                                                                                                              f                              R                2                                      =                          1                              2                ⁢                                                                  ⁢                π                ⁢                                                                                                    L                        tot                                            2                                        ⁢                                          C                      tot                                                                                                          =                                            f                              R                1                                                    f                              R                2                                              =                      2                                              (        3        )            
Although the above-mentioned approach can thus theoretically result in an improvement in the baseband equivalent resonance by a factor of the square root of two, conventional designs employing such an approach have several disadvantages. A first disadvantage associated with conventional designs such as that of FIG. 1 is that implementation of the first and second bias lines (e.g., the bias lines 104, 106) requires an increased printed circuit board (PCB) footprint for proper design implementation.
Additionally, conventional designs also suffer from a second disadvantage in that the equivalent baseband resonance achieved by this type of approach typically is in the range of approximately 160 MHz (Megahertz), which is not sufficient to satisfy current multiband or new 5G LTE (5th generation Long Term Evolution) wideband standards. Indeed, there is an increasing need for base stations—and underlying power amplifiers—that can achieve signal bandwidth (or maximum bandwidth) on the order of 200 to 400 MHz (or even beyond), with accompanying higher power levels and current levels, as well as an increasing desire to implement radios that support multiple channels. The aforementioned conventional designs are inadequate for addressing such performance goals. Relatedly, the traditional high value capacitor (HiC) type of baseband termination generally cannot be effectively used in conjunction with shunt-L pre-match due to the impacts of the second harmonic.
For at least these reasons, therefore, it would be advantageous if one or more improved circuits, systems and methods for achieving enhanced baseband resonance, and/or one or more other advantages, could be developed.